1. Field of Invention
The present invention relates to the measurement of incident and reflected waveforms for microwave and radio-frequency (RF) devices-under-test (DUTs) under realistic large signal operating conditions.
2. Description of the Related Art
Modern wireless telecommunication systems use complex signals at high carrier frequencies, with frequencies typically in the GHz range. These signals are generated by electrical circuitry, like e.g. modulators and mixers that can typically only handle low power levels in the milliwatt range. The generated low power signals are amplified to a higher power level before being sent to the antenna. At the antenna power levels range from about 10 milliwatt for a cellular phone to about 100 Watt for a base station. The amplification of the signals is performed by means of high frequency power amplifiers. These amplifiers contain one or more high frequency power transistors. In order to build a good amplifier the designer needs a detailed knowledge of the behavior of the high frequency power transistors under a wide range of operating conditions. The knowledge of the transistor behavior is gained by using advanced microwave measurement set-ups and methods. The most advanced method that is available in the prior art is called the “time domain loadpull measurement” method, as described in the paper “Measurement and Control of Current/Voltage Waveforms of Microwave Transistors Using a Harmonic Load-Pull System for the Optimum Design of High Efficiency Power Amplifiers,” authored by D. Barataud et al., IEEE Transactions on Instrumentation and Measurement, Vol. 48, no. 4, pp. 835-842, August 1999. With this method the transistor terminals are excited by signals that are very similar to the actual signals that the transistor would experience in an actual power amplifier circuit and the time domain voltage and current waveforms are determined as they appear at the transistor terminals under the given realistic large signal operating conditions. Our invention relates to a novel apparatus and an improved method to characterize a high-frequency device-under-test under such realistic operating conditions.
In the following we will present the prior art, which is illustrated in FIG. 1. First we will explain the prior art of realizing realistic large signal operating conditions. This is achieved by using one or more high frequency signal generators 11 in combination with variable impedance terminations 15, also called tuners. There are two kinds of tuners: active and passive tuners. Passive tuners generate variable impedances by simply reflecting microwave energy on a moveable mechanical structure (U.S. Pat. No. 6,297,649 by Tsironis) or on a controllable passive electronic circuit (U.S. Pat. No. 5,034,708 by Adamian). Note that “passive electronic circuit” refers to the fact that the electronic circuit does not contain any signal amplifier. Active tuners generate a variable impedance not by simply reflecting microwave energy, but by actually generating a signal (U.S. Pat. No. 3,789,301 by Malaviya) or by amplifying (U.S. Pat. No. 6,509,743 by Ferrero) the output signal generated by the transistor terminal and sending it back towards the terminal while controlling the phase and the amplitude of the generated or amplified signal.
Next we will explain the prior art of measuring the voltage and current waveforms at the terminals of the device-under-test. With existing prior art loadpull measurement setups the voltage and current waveforms are derived from the incident and reflected waveforms, which are sensed by means of distributed directional couplers 12 and 13 (U.S. Pat. No. 2,512,191 by Wolf) or by resistive bridges structures (U.S. Pat. No. 5,121,067 by Marsland), whereby the sensed signals are send to a single or multi-channel broadband microwave signal receiver 17. The distributed directional couplers 12 and 13 and resistive bridges are leveraged from existing vector network analyzer technology and are used because of their inherent high directivity.
One of the major difficulties of the prior art loadpull measurement methods is the generation, at the terminals of the transistor 14, of load impedances having a resistive part that is much higher or much lower than the characteristic impedance of the structures that are used to guide the microwave energy towards and away from said terminals. This characteristic impedance is typically 50 Ohms for today's loadpull setups, whereby one often needs to generate impedances with resistive parts as high as 1000 Ohms or as low as 1 Ohm. The difficulty arises because of significant energy losses in the distributed directional couplers 12 and 13 or the resistive bridge structures, which are typically placed in between the transistor 14 output terminal and the tuner 15. The losses in the distributed directional couplers 12 and 13 are called the insertion losses and their main causes are the skin effect losses in the metal that is used to construct the distributed directional coupler waveguiding structure. The insertion loss is significant because a distributed coupler has a length, which typically equals several times the wavelength corresponding to the highest frequency at which the coupler operates, typically more than 100 mm. The insertion losses of a directional resistive bridge are caused by the resistors, which are always present in such a structure.
Two solutions are present in the prior art to overcome the difficulties mentioned above and to generate very low or very high resistive parts for the load impedance. The first solution is to use active tuners, as described in the abovementioned reference by Barataud et al. (1999). The amplifier inside the active tuner can fully compensate for said energy losses. The disadvantage of this solution is that active tuners are expensive and difficult to control across a wide frequency bandwidth when compared to passive tuners. Another disadvantage is that the active tuning approach has a low power handling capability when compared to passive tuners. This limitation is caused by the inevitable saturation of the amplifiers inside the active tuning circuitry. As a result active tuning approaches are seldom used for power levels above 10 Watt. Another solution is illustrated in FIG. 2. The energy losses of the distributed directional coupler 13 are avoided by putting it after the tuner 15, instead of between the tuner 15 and the output terminal of the device-under-test (DUT) 14. The disadvantage of this approach is that it is difficult to get accurate measurements of voltage and current waveforms. The difficulty is caused by the fact that with this solution the tuner 15 is in the path between the signal receiver 17 and the transistor 14 output terminal and as such said tuner 15 introduces distortions that are a function of the impedance state of the tuner 15. Advanced and lengthy calibrations procedures are needed in order to eliminate these distortions and, even with said calibration procedures available, one can never avoid the loss of measurement sensitivity when the tuner 15 reflects substantially all of the energy back towards the output terminal of DUT 14 without sending any relevant energy towards the distributed coupler 13. This drawback is most critical when it is necessary to measure information on the harmonics that are generated by the nonlinear behavior of the DUT 14. This results in noisy measurements, especially when the DUT 14 is saturated by a large amplitude input signal.
The voltage waveforms, current waveforms or RF power levels are measured by means of a single or multi-channel broadband microwave signal receiver 17. These signal receivers are typically connected to the coupled ports of distributed directive couplers 12 and 13, as such they sense a fraction of the energy of the incident as well as the reflected wave that travels through the main line of the distributed directive couplers 12 and 13. Several types of broadband microwave signal receivers are used, each type offering more or less information about the incident and reflected traveling waves going through the distributed directive couplers 12 and 13. The most simple broadband microwave signal receiver is a set of RF power meters, which only give access to the RF power in Watts or dBm. Some very simple setups can avoid distributed directive couplers and make direct use of a RF power meter as an RF load. More complex systems make use of distributed directive couplers and a vector network analyzer (VNA) as a signal receiver. Such a VNA with a proper power reference calibration provides both the amplitude and the phase of all incident and reflected traveling waves at the frequencies of interest. At any given frequency, when the incident and reflected wave amplitudes and phases are known at the output of the coupled arms of the distributed directive couplers 12 and 13, a proper calibration provides the incident and reflected RF voltage waves at the terminals of the DUT 14. Due to the nonlinearities of the DUT, spectral RF components are often generated at a set of harmonic frequencies; these spectral RF components are called harmonics. This is usually the case when large signals are applied to the DUT 14. The amplitudes and phases of these harmonics are of great interest for the knowledge of the DUT 14 electrical response. With said VNA, the fundamental frequency information is available in addition with the information at harmonic frequencies, but the phases of the harmonics relative to the fundamental component are not available. As such it is impossible, when using a VNA as a receiver 17, to reconstruct the time domain waveform by superposing the fundamental and harmonic waveforms. For that reason, a new generation of broadband RF receivers is sometimes employed: the Large Signal Network Analyzer (LSNA). As the main characteristic of large signal measurements of nonlinear devices like RF transistors is the harmonic frequency component generation, the LSNA, employed with a proper calibration, provides the amplitudes and phases of all measured frequencies, including harmonic frequencies, while preserving the phase relationship between all measured spectral components. The LSNA receiver inputs are typically connected to both coupled arms of the distributed directive couplers 12 and 13. Besides the LSNA, two other kinds of broadband RF receivers are sometimes used to measure time domain waveforms at the DUT 14 terminals: a so-called Microwave Transition Analyzer, in essence a sampling frequency convertor, and a high-speed oscilloscope.
3. Object and Advantages of the Present Invention
The present invention, as illustrated in FIG. 6 has the following three advantages, which are not simultaneously present in any system described in the prior art. First the impedance matching capability of the impedance tuning equipment 15 is substantially not affected by the insertion of the loop type coupling structure 19 because of the inherent low loss of said loop type directive coupling structure 19, in contrast to prior art where one uses a relatively high loss distributed coupler 13 or a resistive bridge. A second advantage is that the calibration procedure remains simple because the calibration coefficients do not depend on the variable impedance setting of the impedance tuning equipment 15. This is the case since the directive coupling structure 19 is inserted between the output terminal of the DUT 14 and the impedance tuning equipment 15, in contrast to some prior art, where the impedance tuning equipment 15 is placed in between the coupler 13 and the output terminal of the DUT 14. A third advantage is that there is no accuracy problem when the impedance tuning equipment 15 is set to reflect substantially all of the energy of a voltage wave back towards the terminal of the DUT 14. With such a setting of the impedance tuning equipment substantially no energy is send to the broadband RF receiver 17 in case the impedance tuning equipment 15 is placed between the coupler 13 and the terminal of the DUT 14. This causes noisy measurements with the abovementioned prior art.